Various switching power supply circuits are widely known including, for example, a switching power supply circuit of the flyback converter type or the forward converter type. The switching converters of the types mentioned are restricted in suppression of switching noise because the switching operation waveform is a rectangular waveform. Further, it is known that the switching converters are limited in enhancement in the power conversion efficiency from their operation characteristics.
Therefore, various switching power supply circuits which rely upon various resonance type converters have been proposed by the applicant of the present application. According to the resonance type converters, a high power conversion efficiency can be obtained readily, and low noise is achieved because the switching operation waveform is a sine waveform. Further, the resonance type converters have a merit also that they can be formed from a comparatively small number of parts.
FIG. 21 is a circuit diagram showing an example of a configuration of a power supply circuit which can be configured based on the invention proposed formerly by the applicant of the present application. The power supply circuit adopts a self-excited current resonance type converter.
The switching power supply circuit shown in this figure includes, as a rectification circuit system for producing a DC input voltage (rectified smoothed voltage Ei) from a commercial AC power supply (AC input voltage VAC), a voltage doubler rectification circuit including two rectification diodes D1 and D2 of the low speed recovery type and two smoothing capacitors Ci1 and Ci2 connected in such a manner as seen in the figure. In the voltage doubler rectification circuit, a rectified smoothed voltage Ei corresponding to twice the AC input voltage VAC is generated across the smoothing capacitors Ci1 and Ci2 connected in series.
The switching converter of the power supply circuit includes two switching elements Q1 and Q2 connected in half bridge connection and interposed between a positive electrode side node of the smoothing capacitor Ci1 and the ground point as seen in the figure. In this instance, a bipolar transistor (BJT; junction type transistor) having a withstanding voltage of 400 V is adopted for the switching elements Q1 and Q2.
Starting resistors RS1 and RS2 are inserted between the collector and the base of the switching elements Q1 and Q2, respectively.
Further, clamp diodes DD1 and DD2 are connected between the base and the emitter of the switching elements Q1 and Q2, respectively. In this instance, the cathode of the clamp diode DD1 is connected to the base of the switching element Q1, and the anode of the clamp diode DD1 is connected to the emitter of the switching element Q1. Meanwhile, the cathode of the clamp diode DD2 is connected to the base of the switching element Q2 and the anode of the clamp diode DD2 is connected to the emitter of the switching element Q2 similarly.
A series connection circuit of a base current limiting resistor RB1, a resonating capacitor CB1 and a driving winding NB1 is interposed between the base of the switching element Q1 and the collector of the switching element Q2. The capacitance of the resonating capacitor CB1 itself and the inductance LB1 of the driving winding NB1 cooperatively form a series resonance circuit.
Similarly, another series connection circuit of a base current limiting resistor RB2, a resonating capacitor CB2 and a driving winding NB2 is interposed between the base of the switching element Q2 and the primary side ground, and the resonating capacitor CB2 and the inductance LB2 of the driving winding NB2 cooperatively form a series resonance circuit for self oscillation.
An orthogonal control transformer PRT (Power Regulating Transformer) drives the switching elements Q1 and Q2 and performs constant voltage control in such a manner as hereinafter described.
The orthogonal control transformer PRT is formed as a saturable reactor of the orthogonal type wherein the driving windings NB1 and NB2 and a resonance current detection winding NA for detecting resonance current are wound and a control winding NC is wound in a direction orthogonal to those of the driving windings NB1 and NB2.
The orthogonal control transformer PRT is structured such that, though not shown in the drawings, two double channel-shaped cores having four magnetic legs are joined together at ends of the magnetic legs thereof to form a solid core. The resonance current detection winding NA and the driving winding NB are wound in the same winding direction on predetermined two ones of the magnetic legs of the solid core. Further, the control winding NC is wound in a direction orthogonal to that of the resonance current detection winding NA and the driving winding NB.
In this instance, the driving winding NB1 is connected at one end thereof to the base of the switching element Q1 through a series connection of the resonating capacitor CB1 and the base current limiting resistor RB1 and at the other end thereof to the collector of the switching element Q2. The driving winding NB2 is connected at one end thereof to the ground and at the other end thereof to the base of the switching element Q2 through a series connection of the resonating capacitor CB2 and the resistor RB2. The driving winding NB1 and the driving winding NB2 are wound such that voltages of the opposite polarities to each other are generated therein.
Meanwhile, the resonance current detection winding NA is connected at one end thereof to a node (switching output point) between the emitter of the switching element Q1 and the collector of the switching element Q2 and at the other end thereof to one end of a primary winding N1 of an insulating converter transformer (Power Isolation Transformer) PIT hereinafter described. It is to be noted that the number of turns (turn number) of the resonance current detection winding NA is, for example, approximately 1 T (turn).
The insulating converter transformer PIT transmits a switching output of the switching elements Q1 and Q2 to the secondary side.
The insulating converter transformer PIT is structured such that, as shown in FIG. 19, it includes an E-E type core formed from E type cores CR1 and CR2 made of, for example, a ferrite material and combined in such a manner that the magnetic legs thereof are opposed to each other as shown in FIG. 19. The primary winding N1 (N4) and a secondary winding N2 (N3) are wound in a divided state on the central magnetic leg of the E-E type core making use of a divisional bobbin B. In this instance, a Litz wire of approximately 60 mmφ is wound in a pattern winding on the divisional bobbin B to form the primary winding N1 (N4) and the secondary winding N2 (N3).
Further, in this instance, a gap G of 0.5 mm to 1.0 mm is formed in the central magnetic leg of the E-E type core. By the gap G, the coupling coefficient k of the primary winding N1 and the secondary winding N2 (N3) is set so that a loose coupling state of, for example, k≈0.8 is obtained.
The primary winding N1 of the insulating converter transformer PIT is connected at one end thereof to a node (switching output point) between the emitter of the switching element Q1 and the collector of the switching element Q2 through the resonance current detection winding NA so that a switching output may be obtained. The primary winding N1 is connected at the other end thereof to the primary side ground through a primary side series resonance capacitor C1 formed from, for example, a film capacitor.
In this instance, the primary side series resonance capacitor C1 and the primary winding N1 are connected in series, and the capacitance of the primary side series resonance capacitor C1 and a leakage inductance component of the insulating converter transformer PIT including the primary winding N1 (series resonance winding) cooperatively form a series resonance circuit for making operation of the switching converter as that of the current resonance type.
A primary side partial voltage resonance capacitor Cp for primary side partial voltage resonance is connected in parallel between the collector and the emitter of the switching element Q2, and the primary side partial voltage resonance capacitor Cp and the leakage inductance of the primary winding N1 form a partial voltage resonance circuit. Consequently, the switching elements Q1 and Q2 can perform zero voltage switching (ZVS) operation and zero current switching (ZCS).
With the primary side switching converter shown in the figure, operation of the current resonance type by the primary side series resonance circuit (L1-C1) and partial voltage resonance operation by the partial voltage resonance circuit (Cp//L1) are obtained.
In short, the power supply circuit shown in the figure adopts a form wherein a resonance circuit for making the primary side switching converter that of the resonance type is combined with another resonance circuit. In the present specification, a switching converter of the type just described is referred to as composite resonance type converter.
Further, on the secondary side of the insulating converter transformer PIT shown in the figure, the secondary windings N2 and N3 are wound independently of each other. Further, a bridge rectification circuit DBR and a smoothing capacitor C01 are connected to the secondary winding N2 to produce a secondary side DC output voltage E01. Meanwhile, a center tap is provided for the secondary winding N3, and rectification diodes D01 and D02 and a smoothing capacitor C02 are connected in such a manner as seen in the figure to the secondary winding N3 to form a full wave rectification circuit consisting of the [rectification diodes D01 and D02 and smoothing capacitor C02] to produce a secondary side DC output voltage E02.
In this instance, the secondary side DC output voltage E01 is branched and inputted also to a control circuit 1.
The control circuit 1 supplies DC current, whose level varies, for example, in response to the level of the secondary side DC output voltage E01 on the secondary side, as control current to the control winding NC of the orthogonal control transformer PRT to perform constant voltage control as hereinafter described.
As switching operation of the power supply circuit having the configuration described above, when a commercial AC power supply is made available first, starting current is supplied to the base of the switching elements Q1 and Q2, for example, through the starting resistors RS1 and RS2, and, for example, if the switching element Q1 is turned on first, then the switching element Q2 is controlled so as to be turned off. Then, resonance current flows as an output of the switching element Q1 through the resonance current detection winding NA→primary winding N1→primary side series resonance capacitor C1. The switching elements Q1 and Q2 are controlled so that the switching element Q2 is turned on and the switching element Q1 is turned off in the proximity of a point of time at which the resonance current becomes zero. Then, resonance current in the reverse direction flows through the switching element Q2. Thereafter, self-excited switching operation wherein the switching elements Q1 and Q2 are turned on alternately is started.
Since the switching elements Q1 and Q2 repeat on/off operations alternately using the terminal voltage of the smoothing capacitor Ci as an operation power supply in this manner, drive current having a waveform proximate to a resonance current waveform is supplied to the primary winding N1 of the insulating converter transformer PIT and an alternating output is obtained at the secondary windings N2 and N3.
The constant voltage control by the orthogonal control transformer PRT is performed in the following manner.
For example, if the secondary side DC output voltage E01 is varied by a variation of the AC input voltage or the load power, then the control circuit 1 variably controls the level of the control current to flow to the control winding NC in response to the variation of the secondary side DC output voltage E01.
By an influence of magnetic fluxes which are generated in the orthogonal control transformer PRT by the control current, the state of a saturation tendency of the orthogonal control transformer PRT changes, which acts to vary the inductance of the driving windings NB1 and NB2. As a result, the conditions of the self-excited oscillation circuit are varied to control the switching frequency fs to vary.
In the power supply circuit shown in the figure, the switching frequency fs is set in a frequency region higher than the resonance frequency of the series resonance circuit of the primary side series resonance capacitor C1 and the primary winding N1, and, for example, if the switching frequency fs becomes higher, then the switching frequency fs is spaced away from the resonance frequency of the series resonance circuit. Consequently, the resonance impedance of the primary side series resonance circuit to the switching output becomes higher.
Since the resonance impedance becomes higher in this manner, the drive current to be supplied to the primary winding N1 of the primary side series resonance circuit is suppressed, and as a result, the secondary side DC output voltage is suppressed thereby to achieve constant voltage control.
FIG. 22 is a circuit diagram showing an example of a configuration of another power supply circuit which can be configured based on the invention proposed formerly by the applicant of the present application. It is to be noted that like elements to those of the power supply circuit shown in FIG. 21 are denoted by like reference characters and description thereof is omitted herein.
Also the power supply circuit shown in FIG. 22 includes a current resonance type converter wherein two switching elements Q11 and Q12 are connected in a half bridge connection, but the driving method thereof is the separate excitation method. In this instance, a MOS-FET or an IGBT (Insulated Gate Bipolar Transistor) is adopted for the switching elements Q11 and Q12.
Further, in this instance, a rectification smoothing circuit formed from a bridge rectification circuit Di and a smoothing capacitor Ci rectifies and smoothes an AC input voltage VAC of a commercial AC power supply AC to produce a DC input voltage, for example, equal to a peak value of the AC input voltage VAC.
The switching elements Q11 and Q12 are connected at the gate thereof to an oscillation and drive circuit 11. The drain of the switching element Q11 is connected to the positive electrode of the smoothing capacitor Ci while the source of the switching element Q11 is connected to the primary side ground through the primary winding N1 and the primary side series resonance capacitor C1. Meanwhile, the drain of the switching element Q12 is connected to the source of the switching element Q11, and the source of the switching element Q12 is connected to the primary side ground.
Also here, a primary side partial voltage resonance capacitor Cp for primary side partial voltage resonance is connected in parallel to the drain-source of the switching element Q12.
Further, clamp diodes DD1 and DD2 are connected in parallel between the drain and the source of the switching elements Q11 and Q12.
The switching elements Q11 and Q12 are driven for switching so that switching operation similar to that described hereinabove with reference to FIG. 21 by the oscillation and drive circuit 11 may be obtained.
In particular, the control circuit 1 in this instance supplies current or a voltage of a level, which varies in response to a variation of the secondary side DC output voltage E01, to the oscillation and drive circuit 11 on the primary side through a photo-coupler PC. The oscillation and drive circuit 11 outputs a switching driving signal (voltage), whose period varies in response to the output level from the control circuit 1, alternately to the gates of the switching elements Q11 and Q12 so that stabilization of the secondary side DC output voltage E01 may be achieved. The switching frequency fs of the switching elements Q11 and Q12 is varied thereby.
In this instance, the oscillation and drive circuit 11 inputs a DC voltage E3 of a low voltage obtained by a rectification circuit formed from a rectification diode D3 and a capacitor C3 to the tertiary winding N4 formed on the primary side of the insulating converter transformer PIT so as to use the DC voltage E3 as an operation power supply. Further, the rectified smoothed voltage Ei is inputted through a starting resistor RS to start the oscillation and drive circuit 11.
FIG. 23 is a view showing operation waveforms of principal components of the power supply circuit shown in FIG. 21.
It is to be noted that also the operation waveforms of the power supply circuit shown in FIG. 22 are substantially similar to the operation waveforms of FIG. 23.
First, if the commercial AC power supply is made available and starting current is supplied, for example, through the starting resistor RS1 to the base of the switching element Q1 to turn on the switching element Q1, then the switching element Q2 is controlled so as to be turned off. Then, as an output of the switching element Q1, resonance current flows along the primary winding N1→primary side series resonance capacitor C1, and the switching elements Q1 and Q2 are controlled such that the switching element Q2 is turned on while the switching element Q1 is turned off in the proximity of a point of time when the resonance current becomes equal to zero. Thereafter, the switching elements Q1 and Q2 are controlled so as to be alternately turned on.
Consequently, within a period TON within which the switching element Q2 is on and another period TOFF within which the switching element Q2 is off, the collector-emitter voltage VQ2 of the switching element Q2 has such a waveform as shown in (a) of FIG. 23, and switching current IQ2 having such a waveform as shown in (b) of FIG. 23 flows to the collector of the switching element Q2.
Further, though not shown here, the collector-emitter voltage of the switching element Q1 and the switching current flowing to the switching element Q1 side have waveforms having a phase difference by 180° from the collector-emitter voltage VQ2 and the switching current IQ2 of the switching element Q2, respectively. In short, the switching elements Q1 and Q2 perform switching at timings at which they are turned on/off alternately.
In response to the switching operations of the switching elements Q1 and Q2, the primary side series resonance current I1 flowing to the primary side series resonance capacitor C1 has a sine waveform according to the switching period as shown in (c) of FIG. 23. In short, the primary side series resonance current I1 has a resonant waveform according to the current resonance type. An alternating voltage is excited also in the secondary winding N2 by an alternating voltage generated in accordance with current flowing through the primary winding N1.
Then, in response to the alternating voltage generated in the secondary winding N2 in such a manner as described above, such a waveform as shown in (e) of FIG. 23 is obtained as that of an inter-terminal voltage V2 between the input terminal of the positive electrode side and the input terminal of the negative electrode side of the bridge rectification circuit DBR connected to the secondary winding N2. In short, a waveform which is clamped at an absolute value level of a rectified smoothed voltage E0 within a period within which rectification current flows through the bridge rectification circuit DBR. The period within which the inter-terminal voltage V2 is clamped at the absolute value level of E0 corresponds to a period within which rectification current flows, and from this, it is indicated that, depending upon the inter-terminal voltage V2, the current flowing through the secondary winding N2 exhibits a discontinuous mode.
Further, to the primary side partial voltage resonance capacitor Cp connected in parallel to the switching element Q2, resonance current ICP flows only within a short period within which the switching elements Q1 and Q2 are turned on or turned off as shown in (d) of FIG. 23. In short, partial voltage resonance operation is performed.
Consequently, the switching elements Q1 and Q2 are controlled so as to perform ZVS operation and ZCS operation thereby to achieve reduction of the switching loss of the switching elements Q1 and Q2.
FIG. 24 is a view showing variation characteristics within the period TON of the AC-DC power conversion efficiency (ηAC→DC), switching frequency fs and switching element Q2 when the load power Po of the secondary side DC output voltage E01 various from 0 W to 200 W when the AC input voltage VAC is VAC=100 V.
As shown in this figure, in the power supply circuit shown in FIG. 21, the switching frequency fs is controlled so as to decrease as the load power Po increases. Further, the switching element Q2 is controlled simultaneously such that the period TON within which the switching element Q2 is on increases.
Further, the AC-DC power conversion efficiency (ηAC→DC) in this instance is approximately 91.8% when the load power Po is Po=200 W, but approximately 92.4% when the load power Po is Po=150 W. Thus, the highest efficiency state is obtained when the load power Po is Po=150 W.
It is to be noted that, when to obtain such an operation waveform shown in FIG. 23 and a characteristic as shown in FIG. 24, the constants of part devices shown in FIG. 21 are selected in the following manner.
First, as regards the insulating converter transformer PIT, the primary winding N1=secondary winding N2=45 T are wound. Further, the primary side series resonance capacitor C1=0.056 μF and the primary side partial voltage resonance capacitor Cp=330 pF are selected.
FIG. 25 is a view showing a further circuit example of a switching power supply circuit as a related art which can be configured based on the invention proposed by the applicant of the present application formerly. It is to be noted that like elements to those of the power supply circuits shown in FIGS. 21 and 22 are denoted by like reference characters and description thereof is omitted herein.
In the power supply circuit shown in this figure, a partial voltage resonance circuit is combined with a separately excited current resonance type converter. Further, the power supply circuit adopts a configuration which satisfies conditions for the commercial AC power supply AC=100 V system.
Also in the power supply circuit shown in this figure, a full-wave rectification smoothing circuit is provided as an input rectification circuit similarly as in the power supply circuit shown in FIG. 22.
Further, in the present power supply circuit, in order to drive switching elements Q11 and Q12 for switching, an oscillation, drive and protection circuit 2 formed from, for example, an IC for universal use is provided. The oscillation, drive and protection circuit 2 includes an oscillation circuit, a driving circuit and a protection circuit. The oscillation circuit and the driving circuit apply a drive signal (gate voltage) of a required frequency to the gate of the each switching elements Q11 and Q12. Consequently, the switching elements Q11 and Q12 perform switching operation such that they are turned on/off alternately with a required switching frequency.
Meanwhile, the protection circuit of the oscillation, drive and protection circuit 2 detects, for example, an overcurrent or overvoltage state of the power supply circuit and controls the switching operation of the switching elements Q11 and Q12 so that the circuit may be protected.
The secondary winding N2 and another secondary winding N2A having a smaller turn number than that of the secondary winding N2 are wound on the secondary side of the insulating converter transformer PIT. In the secondary side windings, an alternating voltage is excited in response to the switching output transmitted to the primary winding N1.
The secondary winding N2 is provided with a center tap as shown in the figure and connected at the center tap thereof to the secondary side ground. Further, a full-wave rectification circuit formed from rectification diodes D01 and D02 and a smoothing capacitor C01 is connected to the secondary winding N2 as shown in the figure. Consequently, a rectified smoothed voltage E01 is obtained as a voltage across the smoothing capacitor C01. The rectified smoothed voltage E01 is supplied to a load side not shown and is branched and inputted also as a detection voltage for the control circuit 1 which is described below.
Also the secondary winding N2A is connected at a center tap thereof to the secondary side ground, and another full-wave rectification circuit formed from rectification diodes D03 and D04 and a smoothing capacitor C02 is connected to the secondary winding N2A. Consequently, a secondary side DC output voltage E02 is obtained as a voltage across the smoothing capacitor C02. The secondary side DC output voltage E02 is supplied also as an operation power supply for the control circuit 1.
The control circuit 1 supplies a detection output corresponding to a level variation of the secondary side DC output voltage E01 to the oscillation, drive and protection circuit 2. The oscillation, drive and protection circuit 2 drives the switching elements Q11 and Q12 such that the switching frequency is varied in response to the detection output of the control circuit 1 inputted thereto. Since the switching frequency of the switching elements Q11 and Q12 is varied in this manner, the level of the secondary side DC output voltage is stabilized.
Here, in the power supply circuit having the configuration described above, a voltage of approximately 1.7 V is obtained as a DC voltage E3 to be supplied as an operation power supply of the oscillation, drive and protection circuit 2.
A voltage of approximately 135 V is obtained as the secondary side DC output voltage E01. Then, under this condition, the turn number T of the secondary winding N2 is selected so that 5 V/T may be satisfied, and the secondary winding current flowing to the rectification diodes D01 and D02 exhibits a continuous mode.
FIG. 26 is a waveform diagram illustrating operation of the power supply circuit shown in FIG. 25 within a switching period. Here, operation under the conditions that the AC input voltage VAC=100 V and the load power Po=125 W is illustrated.
In this instance, the collector-emitter voltage VQ2 of the switching element Q12 within a period TON within which the switching element Q12 is on and another period TOFF within which the switching element Q12 is off has such a waveform as shown in (a) of FIG. 26, and collector current IQ2 having such a waveform as shown in (b) of FIG. 26 flows to the collector of the switching element Q12.
Further, the period A within which sawtooth waveform current of the negative polarity by the exciting inductance L1 of the primary winding N1 flows as collector current IQ2 through the clamp diode DD2 becomes a power non-transmission period within which power transmission to the load side is not performed.
In contrast, another period B within which resonance current of the positive polarity by a leakage inductance component L1l of the primary winding N1 and the capacitance of the primary side series resonance capacitor C1 flows as collector current IQ2 is a power transmission period within which power transmission to the load side is performed.
In this instance, since such primary winding current I1 as illustrated in (d) of FIG. 26 flows to the primary side series resonance capacitor C1, a voltage V1 which exhibits the opposite polarities within the period TON and the period TOFF as seen in (c) of FIG. 26 is obtained across the primary winding N1 of the insulating converter transformer PIT.
Further, since such secondary side current ID as shown in (f) of FIG. 26 flows between the center tap of the secondary winding N2 and the secondary side ground, the inter-terminal voltage V2 across the winding starting end side of the secondary winding N2 and the secondary side ground has such a waveform as shown in (e) of FIG. 26.
FIG. 27 is a waveform diagram which illustrates operation in the conditions of the AC input voltage VAC=100 V and the load power Po=25 W within a switching period for comparison with the waveform diagram shown in FIG. 26 within a switching period.
In this instance, the collector-emitter voltage VQ2 of the switching element Q12 within a period TON within which the switching element Q12 is on and another period TOFF within which the switching element Q12 is off has such a waveform as shown in (a) of FIG. 27, and switching current IQ2 having such a waveform as shown in (b) of FIG. 27 flows to the collector of the switching element Q12. In this instance, as can be seen from comparison with the waveform shown in (b) of FIG. 26, the period A exhibits an expansion. From this, it can be recognized that, when the load power Po is Po=25 (when the load is low), the power transmission to the load side decreases and the power conversion efficiency drops.
To the primary side series resonance capacitor C1 in this instance, primary side series resonance current I1 of such a sine waveform as shown in (d) of FIG. 27 flows. Consequently, a voltage V1 of such a waveform as shown in (c) of FIG. 27 is obtained across the primary winding N1 of the insulating converter transformer PIT.
Further, since such secondary side current ID as shown in (f) of FIG. 27 flows between the center tap of the secondary winding N2 and the secondary side ground, the inter-terminal voltage V2 between the winding starting end side of the secondary winding N2 and the secondary side ground has such a waveform as shown in (e) of FIG. 27.
FIG. 28 shows variation characteristics of the AC-DC power conversion efficiency (ηAC→DC), switching frequency fs and switching output current IQ1 and IQ2 with respect to a load power variation of the power supply circuit shown in FIG. 25. Here, the characteristics under the conditions of the AC input voltage VAC=100 V and the load power Po=0 W to 125 W are shown.
In this instance, it can be seen that the AC-DC power conversion efficiency (ηAC→DC) has a tendency that it increases as the load increases. In particular, although the AC-DC power conversion efficiency (ηAC→DC) in this instance is, for example, approximately 92% when the load power Po is Po=125 W, when the load power Po ix Po=50 W, the AC-DC power conversion efficiency (ηAC→DC) drops to approximately 89%, and when the load power Po is Po=25 W, the AC-DC power conversion efficiency (ηAC→DC) further drops to approximately 82.5%. In this instance, the AC input power in the no-load condition is 4.2 W.
Meanwhile, the switching frequency fs has a tendency that it increases proportionally as the load decreases.
Further, the peak values of the switching output current IQ1 and IQ2 when the load power Po is Po=125 W is 3.5 Ap, and the peak values of the switching output current IQ1 and IQ2 when the load power Po is Po=25 W is 3.0 Ap.
According to the configuration of the power supply circuit shown in FIG. 25, when the load power Po is Po=25 W, resonance current of the positive polarity is caused to flow by the leakage inductance component L1l (L1l=42 μH) of the primary winding N1 and the capacitance of the primary side series resonance capacitor C1, and the period B within which power transmission to the load side is performed becomes shorter. Then, the power non-transmission period A within which sawtooth waveform current of the negative polarity is caused to flow by the exciting inductance L1 (L1=165 μH) of the primary winding N1 becomes longer. As a result, with the power supply circuit of the configuration shown in FIG. 25, the AC-DC power conversion efficiency (ηAC→DC) in a low load condition drops.
It is to be noted that the measurement results illustrated in FIGS. 26 to 28 are obtained by selecting part elements of the power supply circuit of FIG. 25 in the following manner.
First, with regard to the insulating converter transformer PIT, the gap G is set to G=1.0 mm to select the coupling coefficient k of k=0.87. Then, the primary winding N1=24 T, secondary winding N2=23 T+23 T, and tertiary winding N4=2 T are wound.
Further, the primary side series resonance capacitor C1=0.068 μF and the primary side partial voltage resonance capacitor Cp=470 μF are selected.
It is to be noted that, as another related art relating to the present invention, for example, the official gazette of Japanese Patent Laid-Open No. Hei 8-066025 can be listed.
Incidentally, with a power supply circuit constructed as a current resonance type converter of the half bridge type so as to obtain a DC input voltage by means of a full-wave rectification circuit, there is a limitation to enhancement of the power conversion efficiency. More particularly, the load power with which a power conversion efficiency of approximately 92% can be assured is approximately 120 W in the maximum. For example, within a range of the load power of 125 W to 150 W in a heavier load condition, the power conversion efficiency is equal to or lower than 92%.
Therefore, if it is tried to obtain a higher power conversion efficiency with a power supply circuit which can cope with, for example, maximum load power of 150 W or more, then the power supply circuit is configured so as to obtain a DC input voltage by means of a voltage doubler rectification circuit as shown in FIG. 21. By the configuration, the power conversion efficiency can be enhanced to approximately 93%. However, in this instance, it is necessary to incorporate two smoothing capacitors in the voltage doubler rectification circuit, and the part cost becomes higher as the voltage withstanding property of the switching elements Q1 and Q2 and the resonance capacitors increases.
Further, as a problem common to the related art power supply circuits described above, particularly the power conversion efficiency drops as the load power decreases.
For example, taking the power supply circuit shown in FIG. 25 as an example, the AC-DC power conversion efficiency (ηAC→DC) is approximately 89% when the load power Po is Po=50 W and approximately 82.5% when the load power Po is Po=25 W. Also in a no-load condition wherein the load power Po is Po=0 W, the AC input power is approximately 4.2 W.
In this manner, for the composite resonance type converters which include a current resonance type converter described above as the related art converters, a higher power conversion efficiency is demanded within a range from a heavy load condition to a light load condition.